Methods and receivers of carrier frequency offset detection

ABSTRACT

A receiver used for an orthogonal frequency-division multiplexing (OFDM) system is provided. A signal processing device receives an OFDM symbol and processes the OFDM symbol according to the OFDM symbol and a carrier frequency offset compensation coefficient to generate a processed signal. The OFDM symbol includes pilots which have been hierarchically modulated and the processed signal includes the processed pilots. A signal analysis device collects the processed pilots of the processed signal and detects carrier frequency offset to generate the carrier frequency offset compensation coefficient to the signal processing device according to the processed pilots and a plurality of target decision bit error rates. A channel detection module detects a channel response of the processed signal according to the processed pilots and compensates the processed signal to generate an output data.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority of Taiwan Patent Application No.098129516 filed on Sep. 2, 2009, the entirety of which is incorporatedby reference herein.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates to carrier frequency offset detection, and moreparticularly, to methods and receivers using the hierarchical modulatedpilots to detect carrier frequency offset, applied in orthogonalfrequency division multiplexing systems.

2. Description of the Related Art

Recently, the application of orthogonal frequency division multiplexingsystem has become the most important wireless communication technology.Data can be easily and efficiently transmitted and received in thewireless communication environment because of the high transmission rateof the orthogonal frequency division multiplexing system. Therefore,orthogonal frequency division multiplexing scheme is applied to, forexample, Digital Audio Broadcasting (DAB), Digital VideoBroadcasting-Terrestrial/Handheld (DVB-T/H), Wireless Fidelity (Wi-Fi)and Worldwide Interoperability for Microwave Access (WiMAX) etc., andthe orthogonal frequency division multiplexing scheme is regarded as the4th-Generation Wireless System.

The data is transmitted by a plurality of sub-carriers which overlap andare orthogonal to each other in the orthogonal frequency divisionmultiplexing system. In addition, duplicated data which is copied fromthe end of the symbol is defined as a cyclic prefix (CP) or a guardinterval (GI), and the purpose of the duplicated data is to protect theorthogonal frequency division multiplexing symbol from inter-symbolinterference (ISI) generated by multi-path fading and reflection in thechannels. The bandwidth used by the orthogonal frequency divisionmultiplexing system is divided into a number of sub-bands, and thesub-bands are only affected by the flat fading. Thus, the receiver onlyneeds one simple equalizer to adjust signal gain and compensate for theflat fading of the channel. The orthogonal frequency divisionmultiplexing system has many advantages such as an advantage againstmulti-path fading, high-efficiency bandwidth, low-complexity equalizerand high transmission rate.

However, the orthogonal frequency division multiplexing system isaffected by the Doppler effect in the high speed movement condition suchas high speed rail. The orthogonal frequency division multiplexingsystem which is regarded as a multi-carrier system is very sensitive tocarrier frequency offset (CFO) caused by Doppler effect. The carrierfrequency offset will destroy the orthogonality between the sub-carriersand generate inter-carrier interference (ICI) between the sub-carrierssuch that the performance of the orthogonal frequency divisionmultiplexing system in the environment of the high speed decreases andthe bit error rate increases. Therefore, how to detect carrier frequencyoffset (CFO) to cancel inter-carrier interference (ICI) between thesub-carriers is the most important subject to realize the orthogonalfrequency division multiplexing system.

BRIEF SUMMARY OF THE INVENTION

One objective of the invention is to provide a receiver used for anorthogonal frequency-division multiplexing (OFDM) system, comprising: asignal processing device receiving an OFDM symbol and processing theOFDM symbol according to the OFDM symbol and a carrier frequency offsetcompensation coefficient to generate a processed signal, wherein theOFDM symbol comprises pilots which have been hierarchically modulatedand the processed signal comprises processed pilots; a signal analysisdevice collecting the processed pilots of the processed signal anddetecting a carrier frequency offset to generate the carrier frequencyoffset compensation coefficient to the signal processing deviceaccording to the processed pilots and a plurality of target decision biterror rates; and a channel detection module detecting a channel responseof the processed signal according to the processed pilots andcompensating the processed signal to generate an output data.

Another objective of the invention is to provide a method of carrierfrequency offset detection used for an orthogonal frequency-divisionmultiplexing (OFDM) system, comprising: modulating a plurality of pilotsin an OFDM symbol hierarchically; transmitting the OFDM symbol, whereinthe OFDM symbol is affected by a carrier frequency offset; processingthe OFDM symbol according to the OFDM symbol and a carrier frequencyoffset compensation coefficient to generate a processed signal by asignal processing device, wherein the processed signal having theprocessed pilots; collecting the processed pilots of the processedsignal; demodulating the processed pilots; and detecting carrierfrequency offset to generate the carrier frequency offset compensationcoefficient to the signal processing device according to the demodulatedprocessed pilots and a plurality of target decision bit error rates.

The advantage and spirit of the invention could be better understood bythe following recitations together with the appended drawings.

BRIEF DESCRIPTION OF DRAWINGS

The invention can be more fully understood by reading the subsequentdetailed description and examples with references made to theaccompanying drawings, wherein:

FIG. 1 is a block diagram illustrating a transmitter 100 for generatingand transmitting an Orthogonal Frequency Division Multiplexing (OFDM)symbol which has pilots which have been hierarchically modulatedaccording to an embodiment of the invention.

FIG. 2-1 is a schematic diagram illustrating a hierarchical 64-QAMconstellation and a number of gray codes corresponding to a hierarchical64-QAM constellation according to an embodiment of the invention.

FIG. 2-2 is a schematic diagram illustrating the hierarchical 64-QAMconstellation corresponding to the FIG. 2-1 according to an embodimentof the invention.

FIG. 2-3 is a schematic diagram illustrating a uniform hierarchical64-QAM constellation if λ₁=2 and λ₂=1.

FIG. 2-4 is a schematic diagram illustrating a uniform hierarchical64-QAM constellation if λ₁=1.9, λ₂=1.1.

FIG. 2-5 is a schematic diagram illustrating a uniform hierarchical64-QAM constellation if λ₁=1.8, λ₂=1.2.

FIG. 2-6 is a schematic diagram illustrating a uniform hierarchical64-QAM constellation if λ₁=1.6, λ₂=1.4.

FIG. 3 is a block diagram illustrating a receiver 200 according to anembodiment of the invention.

FIG. 4-1 is a performance diagram of each hierarchical bit error rateusing a uniform hierarchical 64-QAM constellation according to anembodiment of the invention, wherein λ₁ is 2 and λ₂ is 1 in the uniformhierarchical 64-QAM constellation, and the vertical axis represents biterror rate (BER), the first horizontal axis represents the signal andnoise ratio (SNR), and the second horizontal axis represents thenormalized frequency offset.

FIG. 4-2 is a performance diagram of each hierarchical bit error rateusing a non-uniform hierarchical 64-QAM constellation according to anembodiment of the invention, wherein λ₁ is 1.6 and λ₂ is 1.4 in thenon-uniform hierarchical 64-QAM constellation, and the vertical axisrepresents bit error rate (BER), the first horizontal axis representsthe signal and noise ratio (SNR), and the second horizontal axisrepresents the normalized frequency offset.

FIG. 5 is flow chart showing the detection of the signal and noise ratio(SNR) according to an embodiment of the invention, wherein λ₁ is 1.6 andλ₂ is 1.4 in the non-uniform hierarchical 64-QAM constellation.

FIGS. 6-1, 6-2, 6-3, 6-4 and 6-5 are the performance curves of bit errorrate of each level according to an embodiment of the invention, whereinλ₁ is 1.6 and λ₂ is 1.4 in the non-uniform hierarchical 64-QAMconstellation and the normalized frequency offset is 0.001, 0.01, 0.05,0.1 and 0.15 in the FIGS. 6-1, 6-2, 6-3, 6-4 and 6-5 respectively.

FIG. 7 is flow chart that shows the detection of carrier frequencyoffset according to an embodiment of the invention, wherein λ₁ is 1.6and λ₂ is 1.4 in the non-uniform hierarchical 64-QAM constellation.

FIG. 8 is a block diagram illustrating the pilot signal analysis module204 according to an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

The following description is of the best-contemplated mode of carryingout the invention. This description is made for the purpose ofillustrating the general principles of the invention and should not betaken in a limiting sense. The scope of the invention is best determinedby reference to the appended claims.

FIG. 1 is a block diagram illustrating a transmitter 100 for generatingand transmitting an Orthogonal Frequency Division Multiplexing (OFDM)symbol which has pilots which have been hierarchically modulatedaccording to an embodiment of the invention. The receiver 100 maycomprise a data source 101 for generating and outputting the digitaldata bits DATA to a data signal modulator 102. The data signal modulator102 is configured to modulate the received digital data bits DATA. Forexample, the received digital data bits DATA are modulated to generatethe modulated in-phase/quadrature-phase data signal by using binaryphase shift keying (BPSK), quadrature phase shift keying (QPSK) orquadrature amplitude modulation (QAM) such as 16-QAM, 64-QAM and 128-QAMetc. A serial-to-parallel converter 103 will convert the serialmodulated in-phase/quadrature-phase data signal into the parallelmodulated in-phase/quadrature-phase data signal, and the parallelmodulated in-phase/quadrature-phase data signal passed by theserial-to-parallel 103 is outputted to an inverse fast fourier transform(IFFT) converter 107 which has N-points inverse fast fourier transform(IFFT). At the same time, a pilot signal source 104 generates andoutputs a number of pilot signals PILOT to a hierarchical signalmodulator 105. The hierarchical signal modulator 105 is configured tomodulate the received pilot signals PILOT by using a uniformhierarchical quadrature amplitude modulation (QAM) constellation. Inthis embodiment of the invention, the pilot signals PILOT are modulatedto generate the modulated in-phase/quadrature-phase pilot signals byusing an uniform hierarchical 64-QAM constellation or an uniformhierarchical 256-QAM constellation but not using a general modulationscheme such as BPSK or QPSK etc. The hierarchical signal modulator 105adjusts a distribution of the constellation points in the hierarchical64-QAM constellation according to two hierarchical level distance ratiosand the hierarchical signal modulator 105 outputs the modulatedin-phase/quadrature-phase pilot signals to a serial-to-parallelconverter 106. The parallel modulated in-phase/quadrature-phase pilotsignals passed by the serial-to-parallel 106 are outputted to theinverse fast fourier transform (IFFT) converter 107. The modulatedin-phase/quadrature-phase pilot signals and the modulatedin-phase/quadrature-phase data signal are transformed from the timedomain to the frequency domain by the inverse fast fourier transform(IFFT) converter 107, and a number of symbols are outputted andtransferred from parallel form to serial form by a parallel-to-serialconverter 108. Finally, a guard interval (GI) which is a copied sectionfrom the end of the symbol is added to the number of symbols to generateand transmit the OFDM symbols by a cyclic prefix (CP) insertion device109. Adding the guard interval (GI) to the head of the symbol canmaintain the continuity of the OFDM symbols. For example, the guardinterval (GI) is one half of the OFDM symbols length or the eighteenthof the OFDM symbols length. Adding the guard interval (GI) can protectthe OFDM symbols from inter-symbol interference (ISI) generated bymulti-path fading and reflection in the channels. In the transmittingprocedure, the Doppler effect is generated due to high speed movement. Aparameter affected by Doppler effect is a multiplicator shown in theFIG. 1 and the multiplicator is also shown as the carrier frequencyoffset (CFO) caused by Doppler effect.

FIG. 2-1 is a schematic diagram illustrating a hierarchical 64-QAMconstellation and a number of gray codes corresponding to hierarchical64-QAM constellation according to an embodiment of the invention. Asshown in the FIG. 1, the hierarchical signal modulator 105 modulates thereceived pilot signals to generate the modulated I/Q pilot signals byusing hierarchical modulation. In hierarchical modulation, the modulatedI/Q pilot signals corresponding to the received pilot signals are foundaccording to hierarchical 64-QAM constellation.

There are three hierarchical bits in the hierarchical 64-QAMconstellation shown in the FIG. 2-1. The biggest point shown in the FIG.2-1 is determined by the first hierarchical bits. In other words, thetwo level bits in the first level jointly determine the biggest point asQPSK in FIG. 2-1. By the same way, a middle point shown in the FIG. 2-1is determined by the two level bits of the second level, and a smallestpoint shown in the FIG. 2-1 is determined by the two level bits of thethird level bits. For example, a binary digit sequence 111001 isgenerated by the pilot signal source 104. Using FIG. 2-1 as an example,the constellation points corresponding to the binary digit sequence111001 is obtained by the hierarchical 64-QAM constellation. The biggestpoint at the upper-left corner of the hierarchical 64-QAM constellationis obtained according to the first level which has two level bits 11,the middle point at the lower-right corner of the selected biggest pointis obtained according to the second level which has two level bits 10and the selected biggest point, and the smallest point at upper-rightcorner of the selected middle point is obtained according to the thirdlevel which has two level bits 01 and the selected middle point. Inorder to generate I/Q pilot signals corresponding to the pilot signals,the constellation points corresponding to the sequence 111001 isdetermined in FIG. 2-1 according to the above steps. Demodulation is areverse procedure of the modulation. The received I/Q pilot signal ismapped to the constellation points in the hierarchical 64-QAMconstellation, and then a gray code corresponding to one of theconstellation points is obtained. Thus, the three levels, one of whichhas its own level bits, am obtained according to the gray code.

FIG. 2-2 is a schematic diagram illustrating the hierarchical 64-QAMconstellation as shown in FIG. 2-1 according to an embodiment of theinvention. The parameters d₂′, d₂, d₃ represent the distance between theconstellation points in the hierarchical 64-QAM constellationrespectively, wherein the parameter d₂ represents the x-axis distancebetween the constellation point of the first level (the biggest point)and the constellation point of the second level (the middle point) asshown in the FIG. 2-2, the parameter d₃ represents the x-axis distancebetween the constellation point of the second level (the middle point)and the constellation point of the second level (the smallest point) asshown in the FIG. 2-2 and the parameter d₂′ represents the x-axisdistance between the y-axis and the constellation point of the thirdlevel which is closest to the y-axis (the smallest point) as shown inthe FIG. 2-2. The parameters λ₁ and λ₂ are defined as hierarchical leveldistance ratios, where λ₁=d₂/d₂′ λ₂=d₃/d₂′. The distribution of aplurality of constellation points in the hierarchical 64-QAMconstellation is changed if the two hierarchical level distance ratiosare adjusted. FIG. 2-3 is a schematic diagram illustrating a uniformhierarchical 64-QAM constellation if λ₁=2 and λ₂=1 and FIG. 2-4, FIG.2-5 and FIG. 2-6 are a schematic diagram illustrating a non-uniformhierarchical 64-QAM constellation if λ₁=1.9, λ₂=1.1 (shown as FIG. 2-4),λ₁=1.8, λ₂=1.2 (shown as FIG. 2-5), λ₁=1.6, λ₂=1.4 (shown as FIG. 2-6)respectively. The hierarchical 64-QAM constellation has a plurality ofconstellation points which present a non-uniform distribution accordingFIG. 2-4, FIG. 2-5 and FIG. 2-6. The distribution of the constellationpoints in the hierarchical 64-QAM constellation is controlled by the twohierarchical level distance ratios λ₁ and λ₂. The bit error ratecorresponding to each level in the uniform hierarchical 64-QAMconstellation is not influenced by the carrier frequency offset (CFO).However, in the non-uniform hierarchical 64-QAM constellation, each biterror rate has different characteristics in response to the carrierfrequency offset. Therefore, signal to noise ratio (SNR) is firstdetected by using the non-uniform hierarchical QAM constellation, andthen carrier frequency offset (CFO) caused by high speed movement isdetected.

FIG. 3 is a block diagram illustrating a receiver 200 according to anembodiment of the invention. The receiver 200 is configured forreceiving an OFDM symbol affected by carrier frequency offset (CFO), anddetecting and compensating the carrier frequency offset using thehierarchically modulated pilots. The functional blocks in the receiver200 are the inverse of the functional blocks in the transmitter 100 todemodulate the signals. The receiver 200 in the FIG. 3 comprises asignal processing device 201 for receiving an OFDM symbol and processingthe OFDM symbol according to the OFDM symbol and a carrier frequencyoffset compensation coefficient to generate a processed signal, whereinthe OFDM symbol has pilots which have been hierarchically modulated, andthe processed signal having the processed pilots. The signal processingdevice 201 comprises a cyclic prefix (CP) removal device 2011 forremoving the added guard interval (GI) in the transmitter 100. Aserial-to-parallel converter 2012 is configured to transfer the symbolsfrom serial form to parallel form. The symbols are transformed from thefrequency domain to the time domain by the fast fourier transform (FFT)converter 2013 and the fast fourier transform (FFT) converter 2013outputs the I/Q data signals and the I/Q pilot signals. Aparallel-to-serial converter 2014 is configured to convert the I/Q datasignals and the I/Q pilot signals from parallel form to serial form,wherein the I/Q data signals and the I/Q pilot signals are defined asthe processed signals. The receiver 200 in the FIG. 3 further comprisesa signal analysis device 20. The signal analysis device 20 comprises apilot signal collector 202 for collecting the I/Q pilot signals of theI/Q data signals and the I/Q pilot signals outputted by theparallel-to-serial converter 2014. A hierarchical pilot signaldemodulator 203 for demodulating the I/Q pilot signals in the OFDMsymbols and outputting a number of pilots. Using the hierarchical 64-QAMconstellation as an example, each of the I/Q pilot signals in the OFDMsymbols demodulated by the hierarchical pilot signal demodulator 203 canrepresent six bits, wherein the first level comprises the first andsecond bits, the second level comprises the third and fourth bits, andthe third level comprises the fifth and sixth bits. Furthermore, thelevel bits in each level are inputted to a pilot signal analysis module204. In order to detect the signal and noise ratio (SNR) whichrepresents the channel quality, the pilot signal analysis module 204determines an order of bit error rates corresponding to all levels in acertain signal and noise ratio (SNR) according to the demodulatedprocessed pilots. The signal and noise ratio (SNR) is detected usingreceived signal strength indication (RSSI), which is a conventionalmethod. High SNR and good communication is determined by the pilotsignal analysis module 204 when the signal and noise ratio exceeds thepredetermined value, 20 dB as an example. Further, carrier frequencyoffset is detected by the pilot signal analysis module 204 to generatethe carrier frequency offset compensation coefficient CC. A channeldetection module 205 is configured for detecting channel responsesaccording to the collected I/Q pilot signals and compensating thechannel responses. The I/Q data signals are demodulated into digitaldata bits DATA by the channel detection module 205. The channeldetection module 205 comprises a channel estimator 2051 for detectingchannel responses according to the collected I/Q pilot signals and aone-tap equalizer 2052 for compensating the channel response accordingto the detected channel responses to compensate the effect by thechannel fading in order to prevent the transmitted signals from theterrible distortion. A data signal demodulator 2053 transforms I/Q datasignals to the digital data bits DATA.

FIG. 4-1 is a performance diagram of bit error rate at each level usinga uniform hierarchical 64-QAM constellation according to an embodimentof the invention, wherein λ₁ is 2 and λ₂ is 1 in the uniformhierarchical 64-QAM constellation, and the vertical axis represents biterror rate (BER), the first horizontal axis represents the signal andnoise ratio (SNR), and the second horizontal axis represents thenormalized frequency offset. The bit error rate of each level in theuniform hierarchical 64-QAM constellation does not follow the normalizedfrequency offset change as shown in FIG. 4-1. No matter what thenormalized frequency offset is 0.001, 0.01, 0.05, 0.1 or 0.15, bit errorrates in sequence always are: bit error rate of the first level, biterror rate of the second level, bit error rate of the third level (fromlarger bit error rates to small rates). The error floor of level bit ineach level also does not follow the normalized frequency offset ofchange, wherein the representation of the error floor means that theeffect of bit error rate by ICI which is caused by the normalizedfrequency offset is not large when the signal and noise ratio (SNR) islow and bit error rate affected by the normalized frequency offset doesnot follow the increasing signal and noise ratio and decrease when thesignal and noise ratio (SNR) is high. For example, none of the bit errorrates of the first level, second level or third level decrease inresponse to the increasing signal and noise ratio when the normalizedfrequency offset is 1.5 as shown in the FIG. 4-1. The error floor alsomeans that the bit error rate can not decrease in response to anincrease in the signal and noise ratio.

FIG. 4-2 is a performance diagram of the bit error rate of each level byusing a non-uniform hierarchical 64-QAM constellation according to anembodiment of the invention, wherein λ₁ is 1.6 and λ₂ is 1.4 in thenon-uniform hierarchical 64-QAM constellation, and the vertical axisrepresents bit error rate (BER), the first horizontal axis representsthe signal and noise ratio (SNR), and the second horizontal axisrepresents the normalized frequency offset. However, in the non-uniformhierarchical 64-QAM constellation, bit error rate corresponding to eachlevel is different. In other words, the hierarchical modulation willcause each level to have a different degree of protection. The bit errorrate of the third level will become the highest bit error rate when thenormalized frequency offset is 0 and the signal and noise ratio (SNR) issmaller than 10 dB. The above condition means that the protectionability of the third level is the lowest, protection ability of thesecond level is middle and protection ability of the first level is thehighest (the bit error rate of the first level is the lowest). When thesignal and noise ratio (SNR) is between 10˜17 dB, the bit error ratefrom high to low is: the second level, the third level, and the firstlevel and the protection ability from high to low is: the first level,the third level, and the second level. When the signal and noise ratio(SNR) is larger than 17 dB, the bit error rate from high to low is: thesecond level, the first level, and the third level, and the protectionability from high to low is: the third level, the first level, and thesecond level. When the normalized frequency offset is 0.01, only thesecond level has different protection ability. The bit error rate of thesecond level in the high signal and noise ratio (SNR) has the phenomenonof the error floor as shown in the FIG. 4-2. In other words, the carrierfrequency offset will cause the bit error rate of the second level tonot follow the increasing signal and noise ratio as it decreases. Thebit error rate of the second level is increasing when the normalizedfrequency offset is 0.05. The protection ability of the first level islarger than the protection ability of the third level when thenormalized frequency offset is 0.1 and the signal and noise ratio (SNR)is below 20 dB. However, the protection ability of the first level issmaller than the third level when the signal and noise ratio (SNR) islarger than 20 dB. This characteristic is regarded as the method fordetermining the signal and noise ratio (SNR). The phenomenon of theerror floor happens when the bit error rate of the first level is 4·10⁻⁴and the bit error rate of the third level is 2·10⁻⁵ is higher than thesignal and noise ratio (SNR). Therefore, the error floor in each leveloccurs and the order of the protection ability for each level change ischanged by the carrier frequency offset. The flow chart is designed fordetecting the signal and noise ratio (SNR) in the embodiment of theinvention (as shown in the FIG. 5).

FIG. 5 is a flow chart for detecting the signal and noise ratio (SNR)according to an embodiment of the invention, wherein λ₁ is 1.6 and λ₂ is1.4 in the non-uniform hierarchical 64-QAM constellation. In step 501,the bit error rate in the first level (represented as Nerr1), the biterror rate in the second level (represented as Nerr2) and the bit errorrate in the third level (represented as Nerr3) in sequence are compared.The signal and noise ratio (SNR) between 0˜10 dB shows that thecommunication quality is not good and the signal and noise ratio (SNR)is low (step 503) when the condition of “Nerr3>Nerr2>Nerr1” is sustained(step 502). However, when the condition of “Nerr3>Nerr2>Nerr1” is notsustained in step 502, the procedure goes to step 504. When thecondition of “Nerr2>Nerr3>Nerr1” is sustained (step 504—YES) and thenormalized frequency offset is below 0.11 (in step 505—YES), the signaland noise ratio (SNR) is between 10˜20 dB and the signal and noise ratio(SNR) between 10˜20 dB represents that the communication quality isnormal. Then the procedure goes to step 509 and ends. On the contrary,the signal and noise ratio (SNR) is larger than 20 dB and the signal andnoise ratio (SNR) which is larger than 20 dB shows that communicationquality is good when the normalized frequency offset is not below 0.11(step 505—NO). The procedure goes to step 509 and ends. However, whenthe condition of “Nerr2>Nerr3>Nerr1” is not sustained (step 504—NO), andthe condition of “Nerr2>Nerr1>Nerr3” is sustained (step 506), the signaland noise ratio (SNR) is larger than 20 dB and the signal and noiseratio (SNR) which is larger than 20 dB represents good communicationquality. The procedure goes to step 509 and ends there. The signal andnoise ratio (SNR) is determined to obtain the communication qualityaccording to the different property of bit error rates in three levelsas shown in the FIG. 5.

Before the carrier frequency offset has been detected, it should beunderstood that the carrier frequency offset seriously affects thesystem performance in certain condition. FIGS. 6-1, 6-2, 6-3, 6-4 and6-5 are the performance curves of bit error rate for each levelaccording to an embodiment of the invention, wherein λ₁ is 1.6 and λ₂ is1.4 in the non-uniform hierarchical 64-QAM constellation and thenormalized frequency offset is 0.001, 0.01, 0.05, 0.1 or 0.15 in theFIGS. 6-1, 6-2, 6-3, 6-4 and 6-5.

According to FIGS. 6-1-6-5, the curve of each level corresponding to thesignal and noise ratio (SNR) is obtained under the different normalizedfrequency offsets. When the signal and noise ratio (SNR) is low, the ICIcaused by carrier frequency offset is not the major cause for the biterror rate of each level. However, the bit error rate of each levelaffected by carrier frequency offset can not follow the increasingsignal and noise ratio (SNR) and decrease when the signal and noiseratio (SNR) is high. The error floor thus happens. The signal and noiseratio (SNR), and not ICI, is a major cause of affecting systemperformance when the signal and noise ratio (SNR) is low. On thecontrary, the ICI, and not SNR, is a major cause of affecting systemperformance in the high signal and noise ratio (SNR). Even if the signaland noise ratio (SNR) is further increased, the error rate does notdecrease due to carrier frequency offset. Therefore, detecting thecarrier frequency offset accurately in the high signal and noise ratio(SNR) is more important than in the low signal and noise ratio (SNR).The carrier frequency offset caused by the Doppler effect is detected bythe obtained different protection degree of each level using pilotswhich have been hierarchically modulated in this invention.

It is noted that a plurality of cross points are generated when the biterror rates in sequence of the levels are changed, and the plurality ofcross points are cross positions between a plurality of performancecurves of the hit error rates of the levels in different signal andnoise ratio (SNR) under a condition of a fixed carrier frequency offset.For example, the performance curves of the first and third levels arecrossed at the cross point X13 and the performance curves of the secondand third levels are crossed at the cross point X23 when the normalizedfrequency offset is 0.001 as shown in the FIG. 6-1. In the FIG. 6-2, theperformance curves of the first and third levels are crossed at thecross point Y13 and the performance curves of the second and thirdlevels are crossed at the cross point Y23 when the normalized frequencyoffset is 0.01. In the FIG. 6-3, the performance curves of the secondand third levels are only crossed at the cross point Z23 when thenormalized frequency offset is 0.01. Therefore, the cross points areobtained because the property of the bit error rate in each levelcorresponds to the signal and noise ratio (SNR). As shown in FIG. 6-1,bit error rate of each level in sequence is Nerr3>Nerr2>Nerr1 (fromlarge to small) before the cross point X23 appears. However, the biterror rate of each level in sequence is Nerr2>Nerr3>Nerr1 when the crosspoint X23 appears and the cross point X13 does not appear. The bit errorrate of each level in sequence is Nerr2>Nerr1>Nerr3 when the cross pointX13 appears. It should be noted that the cross points will appear whenthe bit error rates of the levels in sequence are changed. Therefore,using the order of the bit error rates at each level is the same methodas using the characteristic of the cross points to detect the signal andnoise ratio (SNR).

In addition, cross points are associated with carrier frequency offsetaccording to FIG. 6-1-6-5. The number of cross points decreases whencarrier frequency offset increases. Carrier frequency offset will changethe bit error rate of each level in sequence, and the cross points willappear when the bit error rate of each level in sequence is changed.Thus, it can be understand that cross points are associated with carrierfrequency offset.

A plurality of cross lines are generated by connecting the plurality ofcross points under a condition of different carrier frequency offsetssuch as the cross lines CL1 and CL2 as shown in the FIG. 4-2. Theplurality of cross lines are configured for tracking a variable velocityand an acceleration of a relative velocity between a transmitter andreceiver and time trajectory description of Doppler effect.

FIG. 7 is flow chart for detecting the carrier frequency offsetaccording to an embodiment of the invention, wherein λ₁ is 1.6 and λ₂ is1.4 in the non-uniform hierarchical 64-QAM constellation. When thesignal and noise ratio (SNR) detected by the pilot signal analysismodule 204 is high, the effect of the carrier frequency offset isimportant. It is assumed that there are three target decision bit errorrates, Nstd1, Nstd2 and Nstd3 and Nstd1=10⁻⁴, Nstd2=10⁻³ and Nstd3=10⁻²respectively before the pilot signal analysis module 204 detects thecarrier frequency offset. The detected carrier frequency offset is moreaccurate if the number of target decision bit error rates areincreasing. However, three target decision bit error rates, Nstd1, Nstd2and Nstd3 are used as an example herein.

According to the FIG. 6-1-6-5, there is an obvious difference betweenbit error rates of the levels affected by carrier frequency offset underλ₁=1.6 and λ₂=1.4 in the non-uniform hierarchical 64-QAM constellation.The flow chart begins in step 701. The detected normalized frequencyoffset is below 0.01 when the condition of “Nerr2<Nstd3” is sustained(step 702—YES). The flow chart ends. However, the flow chart goes tostep 703 when the condition of “Nerr2<Nstd3” is not sustained in step702—NO. Then, the detected normalized frequency offset is between0.01˜0.05 when the condition of “Nerr1<Nstd1” is sustained (step703—YES). Then the flow chart ends. However, the flow chart goes to step704 when the condition of “Nerr1<Nstd1” is not sustained in step 703—NO.Then, the detected normalized frequency offset is between 0.05˜0.08 whenthe condition of “Nerr3<Nstd2” is sustained (step 704—YES). Then theflow chart ends. However, the flow chart goes to step 705 when thecondition of “Nerr3<Nstd2” is not sustained in step 704—NO. Next, thedetected normalized frequency offset is between 0.08˜0.11 when thecondition of “Nerr3<Nstd3” is sustained (step 705—YES). The flow chartwill end. However, the detected normalized frequency offset is between0.11˜0.15 when any of the above conditions are not conformed in step705—NO.

FIG. 7 mentions the rough carrier frequency offset detection. Carrierfrequency offset is detected by comparing the relationship between thebit error rates of all levels, and the plurality of target decision biterror rates. In fact, bit error rates at each level under a certaincarrier frequency offset are different such that carrier frequencyoffset is detected accurately by using the distance between the biterror rates of the levels. The flow chart in FIG. 7 does not limit themethod of carrier frequency offset detection. Carrier frequency offsetis determined and the flow chart can be adjusted according to the amountof the plurality of target decision bit error rates and the bit errorrates of the levels such as four bit error rates of the four levels inthe hierarchical 128-QAM constellation or five bit error rates of thefive levels in the hierarchical 256-QAM constellation.

FIG. 8 is a block diagram illustrating the pilot signal analysis module204 (as shown in the FIG. 3) according to an embodiment of theinvention. The pilot signal analysis module 204 comprises a parallel toserial converter 2401 for converting the demodulated processed pilotsinto a plurality of levels where one of the plurality of levelscomprises a plurality of level bits and outputting the level bits to aplurality of pilot signal registers 2402. The plurality of pilot signalregisters 2402 are configured for storing level bits of each level. Forexample, the pilot signal register 2402-1 stores the level bits of thefirst level, the pilot signal register 2402-2 stores the level bits ofthe second level and the pilot signal register 2402-3 stores the levelbits of the third level in the hierarchical 64-QAM constellation. Aplurality of pilot signal error estimators 2403 in the pilot signalanalysis module 204 are configured for comparing each stored level bitsin the pilot signal registers 2402 and the known level bits of eachlevel of the pilots to obtain bit error rate of each level underdifferent SNR. In other words, the pilot signal error estimator 2403-1detects the difference between the stored level bits of the first leveland the known level bits in the first level of the pilot. The pilotsignal error estimator 2403-2 detects the difference between the storedlevel bits of the second level and the known level bits in the secondlevel of the pilot. The pilot signal error estimator 2403-3 detects thedifference between the stored level bits of the third level and theknown level bits in the third level of the pilot. A plurality of pilotsignal error registers 2404 are configured for storing the bit errorrate of each level. In other words, pilot signal error register 2404-1stores the bit error rate of the first level, and so on. A pilot signalerror analysis device 2405 is configured for determining the signal andnoise ratio or the channel quality according to bit error rates oflevels in sequence as shown in the flow chart of the FIG. 5. Then thepilot signal error analysis device 2405 detects the rough carrierfrequency offset according to the flow chart of carrier frequency offsetdetection as shown in the FIG. 7 when the signal and noise ratio is highor exceeds a predetermined value. Next, a numerical controlledoscillator (NCO) 2406 detects the fine carrier frequency offset togenerate a carrier frequency offset compensation coefficient CC to thesignal processing device 201.

This invention discloses that the original pilots in the OFDM systembecome hierarchically modulated pilots. In the receiver, carrierfrequency offset caused by high speed is detected and signal and noiseratio (SNR) is determined by using the property of each hierarchical bitcorresponding to the different carrier frequency offset. Thus, receivedsignal strength indications (RSSI) for detecting the signal and noiseratio (SNR) are not needed.

With the examples and explanations given above, the features and spiritof the invention are well described. Those skilled in the art willreadily observe that numerous modifications and alterations of theembodiments may be made while retaining the core concept of theinvention. Accordingly, the above disclosure should be construed aslimited only by the needs and bounds of the appended claims.

What is claimed is:
 1. A receiver used for an orthogonalfrequency-division multiplexing (OFDM) system, comprising: a signalprocessing device receiving an OFDM symbol and processing the OFDMsymbol according to the OFDM symbol and a carrier frequency offsetcompensation coefficient to generate a processed signal, wherein theOFDM symbol comprises pilots which have been hierarchically modulatedand the processed signal comprises processed pilots; a signal analysisdevice collecting the processed pilots of the processed signal anddetecting a carrier frequency offset to generate the carrier frequencyoffset compensation coefficient to the signal processing deviceaccording to the processed pilots and a plurality of target decision biterror rates; and a channel detection module detecting a channel responseof the processed signal according to the processed pilots andcompensating the processed signal to generate an output data, whereinthe signal processing device comprises a pilot signal collectorcollecting the processed pilots of the processed signal, a hierarchicalpilot signal demodulator demodulating the processed pilots, and a pilotsignal analysis module determining a signal and noise ratio according tothe demodulated processed pilots, and detecting the carrier frequencyoffset to generate the carrier frequency offset compensation coefficientto the signal processing device according to the demodulated processedpilots and the target decision bit error rates when the signal and noiseratio exceeds a predetermined value.
 2. The receiver of claim 1, whereinthe pilot signal analysis device comprises: a parallel to serialconverter converting the demodulated processed pilots into a pluralityof levels where one of the plurality of levels comprises a plurality oflevel bits and outputting of the level bits in each of the plurality oflevels; a pilot signal error detector detecting bit error rates of thelevels; and a pilot signal error analysis device determining the signaland noise ratio according to the bit error rates in sequence, anddetecting the carrier frequency offset to generate the carrier frequencyoffset compensation coefficient to the signal processing deviceaccording to the bit error rates and the plurality of target decisionbit error rate in sequence when the signal and noise ratio exceeds apredetermined value.
 3. The receiver of claim 1, wherein the processedpilots are demodulated according to a uniform hierarchical quadratureamplitude modulation (QAM) constellation by the hierarchical pilotsignal demodulator.
 4. The receiver of claim 3, wherein a distributionof a plurality of constellation points in the uniform hierarchicalquadrature amplitude modulation (QAM) constellation is adjustedaccording to a plurality of hierarchical level distance ratios.
 5. Thereceiver of claim 2, wherein the pilot signal analysis module furthercomprises a plurality of pilot signal registers for storing the levelbits in each of the plurality of levels respectively.
 6. The receiver ofclaim 2, wherein the pilot signal analysis module further comprises aplurality of pilot signal error registers for storing the bit errorrates.
 7. The receiver of claim 2, wherein the bit error rate comprisesa first level bit error rate, a second level bit error rate and a thirdlevel bit error rate.
 8. The receiver of claim 7, wherein the signal tonoise ratio is determined according to the first level bit error rate,the second level bit error rate and the third level bit error rate insequence, and the signal to noise ratio exceeding the predeterminedvalue is determined when the second level bit error rate exceeds thefirst level bit error rate and the first level bit error rate exceedsthe third level bit error rate.
 9. The receiver of claim 8, wherein thebit error rates and the plurality of target decision bit error rate insequence are determined when the signal to noise ratio exceeds thepredetermined value, and the plurality of target decision bit errorrates comprises a first target decision bit error rate, a second targetdecision bit error rate and a third target decision bit error rate. 10.The receiver of claim 9, wherein the carrier frequency offsetcompensation coefficient is between 0.001 and 0.01 when the signal tonoise ratio exceeds the predetermined value and the second level biterror rate is smaller than the third target decision bit error rate. 11.The receiver of claim 9, wherein the carrier frequency offsetcompensation coefficient is between 0.01 and 0.05 when the signal tonoise ratio exceeds the predetermined value and the first level biterror rate is smaller than the first target decision bit error rate. 12.The receiver of claim 9, wherein the carrier frequency offsetcompensation coefficient is between 0.05 and 0.08 when the signal tonoise ratio exceeds the predetermined value and the third level biterror rate is smaller than the second target decision bit error rate.13. The receiver of claim 9, wherein the carrier frequency offsetcompensation coefficient is between 0.08 and 0.11 when the signal tonoise ratio exceeds the predetermined value and the third level biterror rate is smaller than the third target decision bit error rate. 14.The receiver of claim 9, wherein the carrier frequency offsetcompensation coefficient is between 0.11 and 0.15 when the signal tonoise ratio exceeds the predetermined value and the third level biterror rate is not smaller than the third target decision bit error rate.15. The receiver of claim 1, wherein the channel detection modulefurther comprises a one-tap equalizer for compensating the channelresponse.
 16. The receiver of claim 1, wherein the signal processingdevice further comprises a Fourier transform converter for transformingthe OFDM symbol from time domain to frequency domain.
 17. The receiverof claim 2, wherein a plurality of cross points are generated when thebit error rates in sequence of the levels are changed, and the pluralityof cross points are cross positions between a plurality of performancecurves of the bit error rates of the levels in different signal andnoise ratio (SNR) under a condition of a fixed carrier frequency offset.18. The receiver of claim 17, wherein a plurality of cross lines aregenerated by connecting the plurality of cross points under a conditionof different carrier frequency offsets and the plurality of cross linesare configured for tracking a variable velocity and an acceleration of arelative velocity between a transmitter and receiver.
 19. A method ofcarrier frequency offset detection used for an orthogonalfrequency-division multiplexing (OFDM) system, comprising: modulating aplurality of pilots in an OFDM symbol hierarchically; transmitting theOFDM symbol, wherein the OFDM symbol is affected by a carrier frequencyoffset; processing the OFDM symbol according to the OFDM symbol and acarrier frequency offset compensation coefficient to generate aprocessed signal by a signal processing device, wherein the processedsignal having processed pilots; collecting the processed pilots of theprocessed signal; detecting a channel response of the processed signalaccording to the processed pilots and compensating the processed signalto generate an output data; demodulating the processed pilots to formdemodulated processed pilots; determining a signal and noise ratioaccording to the demodulated processed pilots; and detecting carrierfrequency offset to generate the carrier frequency offset compensationcoefficient to the signal processing device according to the demodulatedprocessed pilots and a plurality of target decision bit error rates whenthe signal and noise ratio exceeds a predetermined value.
 20. The methodof carrier frequency offset detection of claim 19, further comprising:converting the demodulated processed pilots into a plurality of levelswhere one of the plurality of levels comprises a plurality of levelbits; detecting a plurality of bit error rates of the levels; anddetermining a signal and noise ratio according to the bit error rates,and detecting the carrier frequency offset to generate the carrierfrequency offset compensation coefficient to the signal processingdevice according to the bit error rate and the plurality of targetdecision bit error rate in sequence when the signal and noise ratioexceeds a predetermined value.
 21. The method of carrier frequencyoffset detection of claim 19, wherein the processed pilots aredemodulated according to an uniform hierarchical quadrature amplitudemodulation (QAM) constellation by the hierarchical pilot signaldemodulator.
 22. The method of carrier frequency offset detection ofclaim 20, wherein a distribution of a plurality of constellation pointsin the uniform hierarchical quadrature amplitude modulation (QAM)constellation is adjusted according to a plurality of hierarchical leveldistance ratios.
 23. The method of carrier frequency offset detection ofclaim 20, further comprising: storing the level bits and the bit errorrate.
 24. The method of carrier frequency offset detection of claim 20,wherein the bit error rates comprises a first level bit error rate, asecond level bit error rate and a third level bit error rate.
 25. Themethod of carrier frequency offset detection of claim 24, wherein thesignal to noise ratio is determined according to the first level biterror rate, the second level bit error rate and the third level biterror rate in sequence, and wherein the signal to noise ratio isdetermined so that the signal to noise ratio exceeds the predeterminedvalue when the second level bit error rate is larger than the firstlevel bit error rate and the first level bit error rate is larger thanthe third level bit error rate.
 26. The method of carrier frequencyoffset detection of claim 25, wherein the bit error rates and theplurality of target decision bit error rates in sequence are determinedwhen the signal to noise ratio exceeds the predetermined value, and theplurality of target decision bit error rates comprises a first targetdecision bit error rate, a second target decision bit error rate and athird target decision bit error rate.
 27. The method of carrierfrequency offset detection of claim 26, wherein the carrier frequencyoffset compensation coefficient is between 0.001 and 0.01 when thesignal to noise ratio exceeds the predetermined value and the secondlevel bit error rate is smaller than the third target decision bit errorrate.
 28. The method of carrier frequency offset detection of claim 26,wherein the carrier frequency offset compensation coefficient is between0.01 and 0.05 when the signal to noise ratio exceeds the predeterminedvalue and the first level bit error rate is smaller than the firsttarget decision bit error rate.
 29. The method of carrier frequencyoffset detection of claim 26, wherein the carrier frequency offsetcompensation coefficient is between 0.05 and 0.08 when the signal tonoise ratio exceeds the predetermined value and the third level biterror rate is smaller than the second target decision bit error rate.30. The method of carrier frequency offset detection of claim 26,wherein the carrier frequency offset compensation coefficient is between0.08 and 0.11 when the signal to noise ratio exceeds the predeterminedvalue and the third level bit error rate is smaller than the thirdtarget decision bit error rate.
 31. The method of carrier frequencyoffset detection of claim 26, wherein the carrier frequency offsetcompensation coefficient is between 0.11 and 0.15 when the signal tonoise ratio exceeds the predetermined value and the third level biterror rate is not smaller than the third target decision bit error rate.32. The method of carrier frequency offset detection of claim 19,further comprising detecting a channel response of the processed signalaccording to the processed pilots and compensating the processed signalto generate an output data.
 33. The method of carrier frequency offsetdetection of claim 19, further comprising transforming the OFDM symbolfrom time domain to frequency domain.
 34. The method of carrierfrequency offset detection of claim 20, wherein a plurality of crosspoints are generated when the bit error rates in sequence of the levelsare changed, and the plurality of cross points are cross positionsbetween a plurality of performance curves of the bit error rates of thelevels in different signal and noise ratio (SNR) under a condition of afixed carrier frequency offset.
 35. The method of carrier frequencyoffset detection of claim 34, wherein a plurality of cross lines aregenerated by connecting the plurality of cross points under a conditionof different carrier frequency offsets and the plurality of cross linesare configured for tracking a variable velocity and an acceleration of arelative velocity between a transmitter and receiver.